In the low frequency band, most of filter circuits are fabricated by attaching discrete components such as coils or capacitors. In the high frequency band such as microwave or millimeter wave bands, however, they are usually fabricated with distributed constant type circuits.
FIG. 18 is a perspective view showing a configuration of an edge coupled filter as a representative distributed constant filter. This filter is provided for a micro-strip line that is most common as a distributed constant line. In FIG. 18, this filter includes a substrate 150 formed of an insulator such as alumina ceramic. A ground layer 151 is formed on the entire back surface of substrate 150. Lines 152 and 153 are part of the micro-strip line as a high frequency transmission line and respectively form an input terminal and an output terminal for the filter. Lines 154 and 155 form so-called λ/2 open line resonators. As used herein λ is a wavelength of the electrical signal transmitted through the line, in the frequency in the vicinity of the center frequency of the filter circuit. Generally, micro-strip lines 152 and 153 and λ/2 open line resonators 154 and 155 are collectively patterned with high accuracy on the surface of insulator substrate 150 by means of print or photolithography. Therefore the planar circuit filter having the structure in FIG. 18 is generally known as a filter circuit with low cost and with excellent productivity.
In the following, a distributed constant type filter circuit formed of micro-strip lines as shown in FIG. 18 will mainly be described in the present specification. The effect of the present invention, however, is not limited to such a filter. It can readily be applied to a filter circuit formed of a coplanar line or a semi-concentrated constant type filter with a part of circuit elements replaced with concentrated constant discrete components. Furthermore, in the following, only a planar view of a substrate seen from above will be shown as a view showing the structure of the distributed constant filter, for the sake of brevity, in the present specification.
Equivalent circuits having the structure of FIG. 18 are shown in FIGS. 19A and 19B. In the following, in the present specification, the equivalent circuit is shown in two stages to facilitate understanding. First, FIG. 19A is an equivalent circuit that is represented with great use of distributed constant line in a one-to-one correspondence with the structure of FIG. 18. However, the equivalent circuit including a distributed constant line as shown in FIG. 19A is inconvenient in a later simulation. Assuming that calculation is made using a commercially available high frequency circuit simulator, the calculation results somewhat vary depending products and manufactures and methods of defining parameters are so varied to understand. In the present specification, an equivalent circuit with only concentrated constant in FIG. 19B is illustrated together, and the simulation is mainly performed using the equivalent circuit with only concentrated constant. The two types of equivalent circuits in FIGS. 19A and 19B are equivalent in the vicinity of resonance frequency of the filter. This is because a λ/2 open line resonator is equivalent to an LC parallel resonance circuit having one end grounded in the vicinity of its resonance frequency.
In FIGS. 19A and 19B, an LC parallel resonance circuit 154 including a coil 154a and a capacitor 154b having one end grounded and an LC parallel resonance circuit 155 including a coil 155a and a capacitor 155b having one end grounded correspond to λ/2 open line resonators 154 and 155 in FIG. 18, respectively. Each of coils 154a and 155b has a prescribed inductance L1 and each of capacitors 154b and 155b has a prescribed capacitance C1. This is because, in the resonance frequency, the middle portion of the λ/2 open line resonator is equivalently grounded and the impedance is close to infinity at both open ends. Capacitors 156 and 157 having a capacitance C2 in FIGS. 19A and 19B correspond to electromagnetic field coupling portions 156 and 157 in FIG. 18. In electromagnetic field coupling portions 156 and 157, micro-strip lines 152 and 153 and λ/2 open line resonators 154 and 155 are arranged closely spaced apart from each other at the open ends approximately by λ/4 or less. In such a case, it is known that electromagnetic field coupling occurs based on capacitive coupling. Capacitor 158 having a capacitance C3 in FIGS. 19A and 19B corresponds to an electromagnetic field coupling portion 158 in FIG. 18. In electromagnetic field coupling portion 158, lines 154 and 155 are arranged close to each other at the open ends. In this case, it is known that electromagnetic field coupling occurs based on capacitive coupling.
The present invention aims at a filter circuit for use in extremely high frequency band such as millimeter wave band, in particular. An exemplary equivalent circuit in FIG. 19B that is optimally designed for 60 GHz band is shown. FIGS. 20A and 20B are frequency characteristics graphs of that filter. The passband is designed as 58-61 GHz consistently in the present specification. Assume C1=0.3661 pF, C2=0.0527 pF, C3=0.02884 pF, L1=0.01699 nH. In FIGS. 20A and 20B, the axis of abscissas represents the frequency [GHz] and the axis of ordinates represent the absolute value of S parameters expressed in dB. In FIGS. 20A and 20B, S21 representing the pass characteristics and S11 representing the reflection characteristics are plotted at the same time. FIG. 20A shows the characteristics of the wide band and FIG. 20B shows the characteristics of the vicinity of the passband. As can be seen from FIGS. 20A and 20B, the filter having the structure in FIG. 18 functions as a bandpass filter.
In the following, the graphs in the formats shown in FIGS. 20A and 20B will be used to express the filter characteristics. Furthermore, in the present specification, the operation principle of the filter will be described, as shown in FIGS. 18-20B, by first showing the structure, then showing the equivalent circuit thereof, and finally showing the calculation result of the filter characteristics of the equivalent circuit. It is noted that in a second embodiment of the present invention, the effectiveness of the present invention is validated by showing the measurement result of the filter that was actually prototyped rather than only by calculation results.
Bandpass filters with high steepness are in the greatest demand among the filters. For local filters or image filters for the extremely high frequency band such as the millimeter wave band, filters with high steepness are essential, as the passband is in the close vicinity of the attenuation band. On the contrary, for example the filter of FIG. 18 results in the filter characteristics that are gradual with poor steepness as shown in FIGS. 20A and 20B, without some special scheme. Then, in order to improve the steepness of such a bandpass filter, the design method of creating attenuation poles at the frequencies immediately above and below the passband has been developed.
Some specific structures of such a filter having attenuation poles at the frequencies above and below the passband have already been presented in the academy. Among others, for example, a circuit shown in FIG. 21 has been known as a circuit that achieves satisfactory results in the extremely high frequency band such as the millimeter wave band and has a simple structure for facilitating design (“Low Loss Micromachined Filters For Millimeter-Wave Telecommunication Systems”, Pierre Blondy et al., 1998 IEEE MTT-S Digest, pp. 1181-1184).
In FIG. 21, this filter includes an insulator substrate 161 formed of alumina ceramic or the like, micro-strip lines 162 and 163 formed on the surface thereof, and λ/2 open line resonators 164 and 165. Portions 167-170 enclosed by dotted lines are portions where lines 162-165 are close to each other to cause electromagnetic field coupling. It is noted that attention has to be made to the following two points in referring to the aforementioned reference (MTT-S Digest). First, the aforementioned reference assumes that it is characterized in that low loss can be attained by the micromachine technique. However, this is not essential in the operation principle of the filter, and the operation principle itself is same as the filter in FIG. 21. Second, the aforementioned reference describes both of a filter (two-pole filter) having two λ/2 open line resonators and a filter (four-pole filter) having four λ/2 open line resonators. The effect of the invention will be discussed in vain unless the filters are compared under the same conditions. The discussion in the present specification is consistently based on the filter (two-pole filter) having two λ/2 open line resonators, for the sake of brevity.
FIGS. 22A and 22B are circuit diagrams showing the equivalent circuit of the filter in FIG. 21. FIG. 22A is an equivalent circuit with great use of distributed constant line and FIG. 22B is an equivalent circuit represented with only distributed constant. In FIGS. 22A and 22B, an LC parallel resonance circuit 164 including a coil 164a and a capacitor 164b having one end grounded and an LC parallel resonance circuit 165 including a coil 165a and a capacitor 165b having one end grounded correspond to λ/2 open line resonators 164 and 165 in FIG. 21, respectively. Each of coils 164a and 165a has a prescribed inductance L1 and each of capacitors 164b and 165b has a prescribed capacitance C1. Capacitors 168 and 169 having a capacitance C2 in FIGS. 22A and 22B correspond to electromagnetic field coupling portions 168 and 169 in FIG. 21, respectively. A capacitor 167 having a capacitance C3 in FIGS. 22A and 22B corresponds to an electromagnetic field coupling portion 167 in FIG. 21. The coefficient of mutual induction coupling K of FIGS. 22A and 22B corresponds to an electromagnetic field coupling portion 170 in FIG. 21. In electromagnetic field coupling portion 170, the middle portions of two λ/2 open line resonators 164 and 165, that is, the portions where the current is maximum are arranged to align closely parallel to each other. In this case, it is known that electromagnetic field coupling based on mutual inductive magnetic field coupling occurs.
An exemplary equivalent circuit in FIG. 22B that is optimally designed for 60 GHz band is shown. FIGS. 23A and 23B are graphs showing the frequency characteristics of that filter. Assume that C1=0.3546 pF, C2=0.05981 pF, C3=0.00687 pF, L1=0.01846 nH, K=0.0914. As can be seen from FIG. 23A, the following changes take place as compared with FIG. 20A. Attenuation poles are formed at the frequencies above and below the passband, and the steepness of the filter is increased in the vicinity of these attenuation poles. At the attenuation pole on the lower frequency side (48 GHz), S21 in FIG. 20A is −30 dB whereas S21 in FIG. 23A is −50 dB or below. At the attenuation pole on the higher frequency side (69 GHz), S21 in FIG. 20A is −17 dB whereas S21 in FIG. 23A is −50 dB or below. In other words, in a case of a radio communication device in which a local frequency happens to be positioned at 48 GHz relative to the center frequency of 60 GHz, the filter characteristics in FIGS. 23A and 23B are more advantageous than the filter characteristics of FIGS. 20A and 20B, as it provides more attenuation amount.
The filter of the equivalent circuit in FIG. 22A, 22B is a commonly known circuit configuration and is described in many references. For example, it is described in the second chapter of “Design and Application of Communication Filter Circuit” (edited and authored by Yoshihiro Konishi, Sogo Denshi Shuppan), which is a prominent text book of the high frequency filter technique.